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FEATURES ULTRALOW NOISE PERFORMANCE 2.9 nV/ Hz at 10 kHz 0.38 V p-p, 0.1 Hz to 10 Hz 6.9 fA/ Hz Current Noise at 1 kHz EXCELLENT AC PERFORMANCE 12.5 V/ s Slew Rate 20 MHz Gain Bandwidth Product THD = 0.0002% @ 1 kHz Internally Compensated for Gains of +5 (or -4) or Greater EXCELLENT DC PERFORMANCE 0.5 mV Max Offset Voltage 250 pA Max Input Bias Current 2000 V/mV Min Open Loop Gain Available in Tape and Reel in Accordance with EIA-481A Standard APPLICATIONS Sonar Photodiode and IR Detector Amplifiers Accelerometers Low Noise Preamplifiers High Performance Audio
Ultralow Noise, High Speed, BiFET Op Amp AD745
CONNECTION DIAGRAM 16-Lead SOIC (R) Package
amplifier for high-speed applications demanding low noise and high dc precision. Furthermore, the AD745 does not exhibit an output phase reversal. The AD745 also has excellent dc performance with 250 pA maximum input bias current and 0.5 mV maximum offset voltage. The internal compensation of the AD745 is optimized for higher gains, providing a much higher bandwidth and a faster slew rate. This makes the AD745 especially useful as a preamplifier where low level signals require an amplifier that provides both high amplification and wide bandwidth at these higher gains. The AD745 is available in two performance grades. The AD745J and AD745K are rated over the commercial temperature range of 0C to 70C, and are available in the 16-lead SOIC package.
120 120
PRODUCT DESCRIPTION
The AD745 is an ultralow noise, high-speed, FET input operational amplifier. It offers both the ultralow voltage noise and high speed generally associated with bipolar input op amps and the very low input currents of FET input devices. Its 20 MHz bandwidth and 12.5 V/s slew rate makes the AD745 an ideal
1000 RSOURCE
INPUT NOISE VOLTAGE - nV/ Hz
EO 100 RSOURCE
OP37 AND RESISTOR
100 PHASE 80 60 GAIN 40 20 0
100 80 60 40
AD745 AND RESISTOR OR OP37 AND RESISTOR 10
AD745 AND RESISTOR
20 0 -20 100M
RESISTOR NOISE ONLY 1 100 1k 10k 100k SOURCE RESISTANCE - 1M 10M -20 100 1k 10k 100k 1M FREQUENCY - Hz 10M
Figure 1.
Figure 2.
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Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 (c) Analog Devices, Inc., 2002
PHASE MARGIN - Degrees
OPEN-LOOP GAIN - dB
AD745-SPECIFICATIONS
AD745 ELECTRICAL CHARACTERISTICS
Model Conditions INPUT OFFSET VOLTAGE 1 Initial Offset Initial Offset vs. Temp. vs. Supply (PSRR) vs. Supply (PSRR) INPUT BIAS CURRENT 3 Either Input Either Input @ TMAX Either Input Either Input, VS = 5 V INPUT OFFSET CURRENT Offset Current @ TMAX FREQUENCY RESPONSE Gain BW, Small Signal Full Power Response Slew Rate Settling Time to 0.01% Total Harmonic Distortion4 INPUT IMPEDANCE Differential Common Mode INPUT VOLTAGE RANGE Differential5 Common-Mode Voltage Over Max Operating Range 6 Common-Mode Rejection Ratio INPUT VOLTAGE NOISE Min
(@ +25 C and
AD745J Typ 0.25
15 V dc, unless otherwise noted.)
Max 1.0 1.5 100 98 400 8.8 600 200 150 2.2 Min AD745K Typ 0.1 2 106 105 150 250 5.5 400 125 75 1.1 20 120 12.5 5 0.0002 1 x 1010 20 3 x 1011 18 20 +13.3, -10.7 +12 -10 90 88 102 0.38 5.5 3.6 3.2 2.9 6.9 2000 1800 4000 1200 +13, -12 1.0 10.0 6.0 5.0 4.0 +12 nA pA pA pA nA MHz kHz V/s s % pF pF V V V dB dB V p-p nV/Hz nV/Hz nV/Hz nV/Hz fA/Hz V/mV V/mV V/mV V +13.6, -12.6 +12, -10 +13.8, -13.1 40 15 8 50 V V V mA V V mA Max 0.5 1.0 Unit mV mV V/C dB dB pA
TMIN to TMAX TMIN to TMAX 12 V to 18 V2 TMIN to TMAX VCM = 0 V VCM = 0 V VCM = +10 V VCM = 0 V VCM = 0 V VCM = 0 V G = -4 VO = 20 V p-p G = -4 f = 1 kHz G = -4
90 88
2 96
150
250 30 40
250 30 30
20 120 12.5 5 0.0002 1 x 1010 20 3 x 1011 18 20 +13.3, -10.7 -10
VCM = 10 V TMIN to TMAX 0.1 to 10 Hz f = 10 Hz f = 100 Hz f = 1 kHz f = 10 kHz f = 1 kHz VO = 10 V RLOAD 2 k TMIN to TMAX RLOAD = 600 RLOAD 600 RLOAD 600 TMIN to TMAX RLOAD 2 k Short Circuit
80 78
95 0.38 5.5 3.6 3.2 2.9 6.9
5.0 4.0
INPUT CURRENT NOISE OPEN LOOP GAIN
1000 800
4000 1200
OUTPUT CHARACTERISTICS Voltage
+13, -12 +13.6, -12.6 +12, -10 12 20 +13.8, -13.1 40 15 8
Current POWER SUPPLY Rated Performance Operating Range Quiescent Current TRANSISTOR COUNT
20
4.8 # of Transistors
18 10.0
4.8
18 10.0
50
NOTES 1 Input offset voltage specifications are guaranteed after five minutes of operations at T A = 25C. 2 Test conditions: +VS = 15 V, -VS = 12 V to 18 V and +VS = 12 V to +18 V, -VS = 15 V. 3 Bias current specifications are guaranteed maximum at either input after five minutes of operation at T A = 25C. For higher temperature, the current doubles every 10C. 4 Gain = -4, RL = 2 k, CL = 10 pF. 5 Defined as voltage between inputs, such that neither exceeds 10 V from common. 6 The AD745 does not exhibit an output phase reversal when the negative common-mode limit is exceeded. All min and max specifications are guaranteed. Specifications subject to change without notice.
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AD745
ABSOLUTE MAXIMUM RATINGS 1 ESD SUSCEPTIBILITY
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipation2 SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 W Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VS Output Short-Circuit Duration . . . . . . . . . . . . . . . . Indefinite Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and -VS Storage Temperature Range (R) . . . . . . . . . -65C to +125C Operating Temperature Range AD745J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to 70C Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300C
NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to Absolute Maximum Rating conditions for extended periods may affect device reliability. 2 16-Pin Plastic SOIC Package: JA = 100C/W, JC = 30C/W
An ESD classification per method 3015.6 of MIL-STD-883C has been performed on the AD745, which is a class 1 device. Using an IMCS 5000 automated ESD tester, the two null pins will pass at voltages up to 1,000 volts, while all other pins will pass at voltages exceeding 2,500 volts.
ORDERING GUIDE
Model AD745JR-16 AD745KR-16
*
Temperature Range 0C to 70C 0C to 70C
Package Option* R-16 R-16
R = Small Outline IC.
CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD745 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
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AD745 -Typical Performance Characteristics
20 RLOAD = 10k
(@ + 25 C, VS =
15 V, unless otherwise noted.)
35
20 RLOAD = 10k
OUTPUT VOLTAGE SWING - V p-p
30 25 20 15 10 5 0 10
INPUT VOLTAGE SWING - V
15 +VIN 10 -VIN 5
INPUT VOLTAGE SWING - V
15 POSITIVE SUPPLY 10 NEGATIVE SUPPLY 5
0 0 5 10 SUPPLY VOLTAGE 15 VOLTS 20
0 0 5 10 SUPPLY VOLTAGE 15 VOLTS 20
100 1k LOAD RESISTANCE -
10k
TPC 1. Input Voltage Swing vs. Supply Voltage
TPC 2. Output Voltage Swing vs. Supply Voltage
TPC 3. Output Voltage Swing vs. Load Resistance
12
10-6
INPUT BIAS CURRENT - Amps
200 100
QUIESCENT CURRENT - mA
10-7 OUTPUT IMPEDANCE - 10-8 10-9 10-10 10-11 10-12 -60 -40 -20 0.01 10k 10
9
6
1 CLOSED LOOP GAIN = -5 0.1
3
0 0 5 10 SUPPLY VOLTAGE 15 VOLTS 20
0 20 40 60 80 100 120 140 TEMPERATURE - C
100k
1M 10M FREQUENCY - Hz
100M
TPC 4. Quiescent Current vs. Supply Voltage
TPC 5. Input Bias Current vs. Temperature
TPC 6. Output Impedance vs. Frequency
300
10-6
28
GAIN BANDWIDTH PRODUCT - MHz
0 20 40 60 80 100 120 140 TEMPERATURE - C
INPUT BIAS CURRENT - Amps
INPUT BIAS CURRENT - pA
10-7 10-8 10-9 10-10 10-11 10-12 -60 -40 -20
26 24 22 20 18 16 14 -60 -40 -20
200
100
0 -12
-9
-6 -3 0 3 6 9 COMMON-MODE VOLTAGE - V
12
0 20 40 60 80 100 120 140 TEMPERATURE - C
TPC 7. Input Bias Current vs. Common-Mode Voltage
TPC 8. Short Circuit Current Limit vs. Temperature
TPC 9. Gain Bandwidth Product vs. Temperature
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AD745
120 100
OPEN-LOOP GAIN - dB SLEW RATE - V/ s
14
150 RL = 2k
80 60 40 20 0 -20 100 GAIN
12
OPEN-LOOP GAIN - dB
PHASE
140
130
CLOSED-LOOP GAIN = 5 10
120
100
1k
10k 100k 1M FREQUENCY - Hz
10M
100M
8 -60 -40 -20 0 20 40 60 80 100 110 120 TEMPERATURE - C
80
0
10 5 SUPPLY VOLTAGE
15 VOLTS
20
TPC 10. Open-Loop Gain and Phase vs. Frequency
TPC 11. Slew Rate vs. Temperature
TPC 12. Open-Loop Gain vs. Supply Voltage
120 COMMON-MODE REJECTION - dB
120
35 RL = 2k
POWER SUPPLY REJECTION - dB
110 100 90 80 Vcm = 70 60 50 100 10V
OUTPUT VOLTAGE SWING - V p-p
100 +SUPPLY 80
30 25 20 15 10 5 0 10k
60 -SUPPLY 40
20 0 100
1k
10k 100k FREQUENCY - Hz
1M
10M
1k
10k 100k 1M FREQUENCY - Hz
10M
100M
100k 1M FREQUENCY - Hz
10M
TPC 13. Common-Mode Rejection vs. Frequency
TPC 14. Power Supply Rejection vs. Frequency
TPC 15. Large Signal Frequency Response
-60
0.1
NOISE VOLTAGE (referred to input) - nV/ Hz
TOTAL HARMONIC DISTORTION (THD) - dB
TOTAL HARMONIC DISTORTION (THD) - %
-40
1.0
CURRENT NOISE SPECTRAL DENSITY - fA/ Hz
100
1k
10 CLOSED-LOOP GAIN = 5
100
-80 GAIN = +10 -100 GAIN = +100 -120 GAIN = -4
0.01
0.001
1.0
10
0.0001
-140 10
100
1k 10k FREQUENCY - Hz
0.00001 100k
0.1 10
1.0 1 10 100 1k FREQUENCY - Hz 10k 100k
100
1k 10k 100k FREQUENCY - Hz
1M
10M
TPC 16. Total Harmonic Distortion vs. Frequency
TPC 17. Input Noise Voltage Spectral Density
TPC 18. Input Noise Current Spectral Density
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AD745
72 66 60 54 NUMBER OF UNITS
NUMBER OF UNITS
648 TOTAL UNITS = 760 594 540 486 432 378 324 270 216 162 108 54 -5 0 5 10 -10 15 INPUT OFFSET VOLTAGE DRIFT - V/ C 0 2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4 INPUT VOLTAGE NOISE @ 10kHz - nV Hz TOTAL UNITS = 4100
48 42 36 30 24 18 12 6 0 -15
TPC 19. Distribution of Offset Voltage Drift. TA = 25C to 125C
TPC 20. Typical Input Noise Voltage Distribution @ 10 kHz
TPC 21. Offset Null Configuration, 16-Lead Package Pinout
2s
100 90 100 90
500ns
10 0%
10 0%
5V
50mV
TPC 22a. Gain of 5 Follower, 16-Lead Package Pinout
TPC 22b. Gain of 5 Follower Large Signal Pulse Response
TPC 22c. Gain of 5 Follower Small Signal Pulse Response
2s
100 90 100 90
500ns
10 0%
10 0%
5V
50mV
TPC 23a. Gain of 4 Inverter, 16-Lead Package Pinout
TPC 23b. Gain of 4 Inverter Large Signal Pulse Response
TPC 23c. Gain of 4 Inverter Small Signal Pulse Response
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AD745
OP AMP PERFORMANCE JFET VERSUS BIPOLAR
The AD745 offers the low input voltage noise of an industry standard bipolar opamp without its inherent input current errors. This is demonstrated in Figure 3, which compares input voltage noise vs. input source resistance of the OP37 and the AD745 opamps. From this figure, it is clear that at high source impedance the low current noise of the AD745 also provides lower total noise. It is also important to note that with the AD745 this noise reduction extends all the way down to low source impedances. The lower dc current errors of the AD745 also reduce errors due to offset and drift at high source impedances (Figure 4). The internal compensation of the AD745 is optimized for higher gains, providing a much higher bandwidth and a faster slew rate. This makes the AD745 especially useful as a preamplifier, where low-level signals require an amplifier that provides both high amplification and wide bandwidth at these higher gains.
1000 RSOURCE
INPUT NOISE VOLTAGE - nV/ Hz
The 0.1 Hz to 10 Hz noise is typically 0.38 V p-p. The user should pay careful attention to several design details to optimize low frequency noise performance. Random air currents can generate varying thermocouple voltages that appear as low frequency noise. Therefore, sensitive circuitry should be well shielded from air flow. Keeping absolute chip temperature low also reduces low frequency noise in two ways: first, the low frequency noise is strongly dependent on the ambient temperature and increases above 25C. Second, since the gradient of temperature from the IC package to ambient is greater, the noise generated by random air currents, as previously mentioned, will be larger in magnitude. Chip temperature can be reduced both by operation at reduced supply voltages and by the use of a suitable clip-on heat sink, if possible. Low frequency current noise can be computed from the magnitude of the dc bias current ~ = 2qI f B In and increases below approximately 100 Hz with a 1/f power spectral density. For the AD745 the typical value of current noise is 6.9 fA/Hz at 1 kHz. Using the formula:
EO 100 RSOURCE
OP37 AND RESISTOR
AD745 AND RESISTOR OR OP37 AND RESISTOR 10
AD745 AND RESISTOR
= 4kT/R f to compute the Johnson noise of a resistor, expressed as a current, one can see that the current noise of the AD745 is equivalent to that of a 3.45 x 108 source resistance.
n
~ I
RESISTOR NOISE ONLY 1 100 1k 10k 100k SOURCE RESISTANCE - 1M 10M
At high frequencies, the current noise of a FET increases proportionately to frequency. This noise is due to the "real" part of the gate input impedance, which decreases with frequency. This noise component usually is not important, since the voltage noise of the amplifier impressed upon its input capacitance is an apparent current noise of approximately the same magnitude. In any FET input amplifier, the current noise of the internal bias circuitry can be coupled externally via the gate-to-source capacitances and appears as input current noise. This noise is totally correlated at the inputs, so source impedance matching will tend to cancel out its effect. Both input resistance and input capacitance should be balanced whenever dealing with source capacitances of less than 300 pF in value.
LOW NOISE CHARGE AMPLIFIERS
Figure 3. Total Input Noise Spectral Density @ 1 kHz vs. Source Resistance
100
INPUT OFFSET VOLTAGE - mV
OP37G 10
1.0 AD745 KN
As stated, the AD745 provides both low voltage and low current noise. This combination makes this device particularly suitable in applications requiring very high charge sensitivity, such as capacitive accelerometers and hydrophones. When dealing with a high source capacitance, it is useful to consider the total input charge uncertainty as a measure of system noise.
1M 10M
0.1 100
1k
10k 100k SOURCE RESISTANCE -
Charge (Q) is related to voltage and current by the simply stated fundamental relationships: Q = CV and I = dQ dt
Figure 4. Input Offset Voltage vs. Source Resistance
DESIGNING CIRCUITS FOR LOW NOISE
An opamp's input voltage noise performance is typically divided into two regions: flatband and low frequency noise. The AD745 offers excellent performance with respect to both. The figure of 2.9 nV/ Hz @ 10 kHz is excellent for a JFET input amplifier.
As shown, voltage, current and charge noise can all be directly related. The change in open circuit voltage (V) on a capacitor will equal the combination of the change in charge (Q/C) and the change in capacitance with a built-in charge (Q/C).
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AD745
DECIBELS REFERENCED TO 1V/ Hz
Figures 5 and 6 show two ways to buffer and amplify the output of a charge output transducer. Both require the use of an amplifier that has a very high input impedance, such as the AD745. Figure 5 shows a model of a charge amplifier circuit. Here, amplification depends on the principle of conservation of charge at the input of amplifier A1, which requires that the charge on capacitor CS be transferred to capacitor CF, thus yielding an output voltage of Q/CF. The amplifiers input voltage noise will appear at the output amplified by the noise gain (1 + (CS/CF)) of the circuit.
CF RS R1 R2
-100 -110 -120 -130 -140 -150 -160 -170 -180 -190 -200 -210 -220 0.01 0.1 1 10 100 1k 10k NOISE DUE TO RB ALONE NOISE DUE TO IB ALONE 100k FREQUENCY - Hz TOTAL OUTPUT NOISE
CS CB* RB*
A1 R1 CS = R2 CF
Figure 7. Noise at the Outputs of the Circuits of Figures 5 and 6. Gain = 10, CS = 3000 pF, RB = 22 M
Figure 5. A Charge Amplifier Circuit
R1 CB*
However, this does not change the noise contribution of RB which, in this example, dominates at low frequencies. The graph of Figure 8 shows how to select an RB large enough to minimize this resistor's contribution to overall circuit noise. When the equivalent current noise of RB (( 4 kT)/R) equals the noise of I B 2qI B , there is diminishing return in making RB larger.
(
)
5.2
R2 RB* CS RB A2
1010
5.2
RESISTANCE IN
109
*OPTIONAL, SEE TEXT.
Figure 6. Model for A High Z Follower with Gain
5.2
108
The second circuit, Figure 6, is simply a high impedance follower with gain. Here the noise gain (1 + (R1/R2)) is the same as the gain from the transducer to the output. Resistor RB, in both circuits, is required as a dc bias current return. There are three important sources of noise in these circuits. Amplifiers A1 and A2 contribute both voltage and current noise, while resistor RB contributes a current noise of:
N=
5.2
107
5.2
106 1pA
10pA
100pA 1nA INPUT BIAS CURRENT
10nA
~
4k
T f RB
Figure 8. Graph of Resistance vs. Input Bias Current Where the Equivalent Noise 4 kT/R, Equals the Noise of the Bias Current I B 2qI B
where: k = Boltzman's Constant = 1.381 x 10-23 Joules/Kelvin T = Absolute Temperature, Kelvin (0C = 273.2 Kelvin) f = Bandwidth - in Hz (Assuming an Ideal "Brick Wall" Filter) This must be root-sum-squared with the amplifier's own current noise. Figure 5 shows that these two circuits have an identical frequency response and the same noise performance (provided that CS/CF = R1/ R2). One feature of the first circuit is that a "T" network is used to increase the effective resistance of RB and improve the low frequency cutoff point by the same factor.
(
)
To maximize dc performance over temperature, the source resistances should be balanced on each input of the amplifier. This is represented by the optional resistor RB in Figures 5 and 6. As previously mentioned, for best noise performance care should be taken to also balance the source capacitance designated by CB The value for CB in Figure 5 would be equal to CS in Figure 6. At values of CB over 300 pF, there is a diminishing impact on noise; capacitor CB can then be simply a large mylar bypass capacitor of 0.01 F or greater.
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AD745
HOW CHIP PACKAGE TYPE AND POWER DISSIPATION AFFECT INPUT BIAS CURRENT
INPUT BIAS CURRENT - Amps
300 TA = 25 C
As with all JFET input amplifiers, the input bias current of the AD745 is a direct function of device junction temperature, IB approximately doubling every 10C. Figure 9 shows the relationship between bias current and junction temperature for the AD745. This graph shows that lowering the junction temperature will dramatically improve IB.
10-6 VS = 15V TA = 25 C
200
JA = 165 C/W
100
JA = 115 C/W
INPUT BIAS CURRENT - Amps
10-7
JA = 0 C/W
10-8
0
5
10 SUPPLY VOLTAGE -
15 Volts
10-9
Figure 11. Input Bias Current vs. Supply Voltage for Various Values of JA
TJ
10-10
10-11
A
10-12 -60
(J TO DIE MOUNT)
-40 -20 0 20 40 60 80 100 JUNCTION TEMPERATURE - C 120 140
B
Figure 9. Input Bias Current vs. Junction Temperature
TA CASE
(DIE MOUNT TO CASE)
The dc thermal properties of an IC can be closely approximated by using the simple model of Figure 10 where current represents power dissipation, voltage represents temperature, and resistors represent thermal resistance ( in C/watt).
TJ
JC CA
A+
B = JC
Figure 12. Breakdown of Various Package Thermal Resistance
REDUCED POWER SUPPLY OPERATION FOR LOWER IB
PIN
JA
TA
WHERE: PIN = DEVICE DISSIPATION TA = AMBIENT TEMPERATURE TJ = JUNCTION TEMPERATURE JC = THERMAL RESISTANCE - JUNCTION TO CASE CA = THERMAL RESISTANCE - CASE TO AMBIENT
Figure 10. Device Thermal Model
Reduced power supply operation lowers IB in two ways: first, by lowering both the total power dissipation and, second, by reducing the basic gate-to-junction leakage (Figure 11). Figure 13 shows a 40 dB gain piezoelectric transducer amplifier, which operates without an ac coupling capacitor, over the -40C to +85C temperature range. If the optional coupling capacitor, C1, is used, this circuit will operate over the entire -55C to +125C temperature range.
100 C1* 108 ** CT** +5V 10k
From this model TJ = TA+JA PIN. Therefore, IB can be determined in a particular application by using Figure 9 together with the published data for JA and power dissipation. The user can modify JA by use of an appropriate clip-on heat sink such as the Aavid #5801. Figure 11 shows bias current versus supply voltage with JA as the third variable. This graph can be used to predict bias current after JA has been computed. Again bias current will double for every 10C.
TRANSDUCER CT 108
AD745
-5V
*OPTIONAL DC BLOCKING CAPACITOR **OPTIONAL, SEE TEXT
Figure 13. A Piezoelectric Transducer
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AD745
TWO HIGH PERFORMANCE ACCELEROMETER AMPLIFIERS
Two of the most popular charge-out transducers are hydrophones and accelerometers. Precision accelerometers are typically calibrated for a charge output (pC/g).* Figures 14 and 15 show two ways in which to configure the AD745 as a low noise charge amplifier for use with a wide variety of piezoelectric accelerometers. The input sensitivity of these circuits will be determined by the value of capacitor C1 and is equal to: V OUT = QOUT C1
low frequency performance, the time constant of the servo loop (R4C2 = R5C3) should be: R2 C1 Time Constant 10 R1 1+ R3
A LOW NOISE HYDROPHONE AMPLIFIER
The ratio of capacitor C1 to the internal capacitance (CT) of the transducer determines the noise gain of this circuit (1 + CT/C1). The amplifiers voltage noise will appear at its output amplified by this amount. The low frequency bandwidth of these circuits will be dependent on the value of resistor R1. If a "T" network is used, the effective value is: R1 (1 + R2/R3).
*pC
= Picocoulombs g = Earth's Gravitational Constant
C1 1250pF R1 110M (5 22M )
Hydrophones are usually calibrated in the voltage-out mode. The circuit of Figures 16 can be used to amplify the output of a typical hydrophone. If the optional ac coupling capacitor CC is used, the circuit will have a low frequency cutoff determined by an RC time constant equal to: 1 Time Constant 10 R1 2 x CC x 100 where the dc gain is 1 and the gain above the low frequency cutoff (1/(2 CC(100 ))) is equal to (1 + R2/R3). The circuit of Figure 17 uses a dc servo loop to keep the dc output at 0 V and to maintain full dynamic range for IB's up to 100 nA. The time constant of R7 and C1 should be larger than that of R1 and CT for a smooth low frequency response.
R2 1900 R3 100
R4*
C1*
R2 9k R3 1k
CC B AND K TYPE 8100 HYDROPHONE R1 108
AD745
OUTPUT
CT
INPUT SENSITIVITY = -179dB RE. 1V/mPa**
B AND K 4370 OR EQUIVALENT
AD745
OUTPUT 0.8mV/pC
*OPTIONAL DC BLOCKING CAPACITOR **OPTIONAL, SEE TEXT
Figure 16. A Low Noise Hydrophone Amplifier Figure 14. A Basic Accelerometer Circuit
C1 1250pF R1 110M 22M ) R3 1k C2 2.2 F R4 18M AD711 R5 18M C3 2.2 F B AND K 4370 OR EQUIVALENT
The transducer shown has a source capacitance of 7500 pF. For smaller transducer capacitances (300 pF), lowest noise can be achieved by adding a parallel RC network (R4 = R1, C1 = CT) in series with the inverting input of the AD745.
R2 1900 R3 100 R4* 108
(5
R2 9k
C1* R4 16M
OUTPUT
AD745
C2 0.27 F R1 108 CT
AD745
OUTPUT 0.8mV/pC
R5 100k AD711K R6 1M 16M
Figure 15. An Accelerometer Circuit Employing a DC Servo Amplifier
DC OUTPUT 1mV FOR IB (AD745) *OPTIONAL, SEE TEXT
100nA
A dc servo loop (Figure 15) can be used to assure a dc output <10 mV, without the need for a large compensating resistor when dealing with bias currents as large as 100 nA. For optimal
Figure 17. A Hydrophone Amplifier Incorporating a DC Servo Loop
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AD745
DESIGN CONSIDERATIONS FOR I-TO-V CONVERTERS
There are some simple rules of thumb when designing an I-V converter where there is significant source capacitance (as with a photodiode) and bandwidth needs to be optimized. Consider the circuit of Figure 18. The high frequency noise gain (1 + CS/CL) is usually greater than five, so the AD745, with its higher slew rate and bandwidth is ideally suited to this application. Here both the low current and low voltage noise of the AD745 can be taken advantage of, since it is desirable in some instances to have a large RF (which increases sensitivity to input current noise) and, at the same time, operate the amplifier at high noise gain.
RF INPUT SOURCE: PHOTO DIODE, ACCELEROMETER, ECT. CL
1F + +12V 0.01 F -12V 0.01 F +12V 1 2 3 4 5 DIGITAL INPUTS 6 3k 7 -12V 8 0.01 F TOP VIEW 10 9 -12V 2000pF 100pF 16 0.01 F
AD1862
20-BIT D/A CONVERTER
15 +12V 14 13 12 11 10 F + ANALOG COMMON 0.1 F OUTPUT 3 POLE LOW PASS FILTER
AD745
0.1 F
DIGITAL COMMON
IS
RB
CS
AD745
Figure 19. A High Performance Audio DAC Circuit Figure 18. A Model for an l-to-V Converter
In this circuit, the RF CS time constant limits the practical bandwidth over which flat response can be obtained, in fact: fB where: fB = signal bandwidth fC = gain bandwidth product of the amplifier With CL 1/(2 RF CS) the net response can be adjusted to a provide a two pole system with optimal flatness that has a corner frequency of fB. Capacitor CL adjusts the damping of the circuit's response. Note that bandwidth and sensitivity are directly traded off against each other via the selection of RF. For example, a photodiode with CS = 300 pF and RF = 100 k will have a maximum bandwidth of 360 kHz when capacitor CL 4.5 pF. Conversely, if only a 100 kHz bandwidth were required, then the maximum value of RF would be 360 k and that of capacitor CL still 4.5 pF. In either case, the AD745 provides impedance transformation, the effective transresistance, i.e., the I/V conversion gain, may be augmented with further gain. A wideband low noise amplifier such as the AD829 is recommended in this application. This principle can also be used to apply the AD745 in a high performance audio application. Figure 19 shows that an I-V converter of a high performance DAC, here the AD1862, can be designed to take advantage of the low voltage noise of the AD745 (2.9 nV/ Hz) as well as the high slew rate and bandwidth provided by decompensation. This circuit, with component values shown, has a 12 dB/octave rolloff at 728 kHz, with a passband ripple of less than 0.001 dB and a phase deviation of less than 2 degrees @ 20 kHz. fC 2 RF CS
An important feature of this circuit is that high frequency energy, such as clock feedthrough, is shunted to common via a high quality capacitor and not the output stage of the amplifier, greatly reducing the error signal at the input of the amplifier and subsequent opportunities for intermodulation distortions.
40
RTI NOISE VOLTAGE - nV/ Hz
30
20 UNBALANCED
10 BALANCED 2.9nV/ Hz 0 10 100 INPUT CAPACITANCE - pF 1k
Figure 20. RTI Noise Voltage vs. Input Capacitance
BALANCING SOURCE IMPEDANCES
As mentioned previously, it is good practice to balance the source impedances (both resistive and reactive) as seen by the inputs of the AD745. Balancing the resistive components will optimize dc performance over temperature because balancing will mitigate the effects of any bias current errors. Balancing input capacitance will minimize ac response errors due to the amplifier's input capacitance and, as shown in Figure 20, noise performance will be optimized. Figure 21 shows the required external components for noninverting (A) and inverting (B) configurations.
REV. D
-11-
AD745
R1 CB = CS RB = RS FOR RS >> R1 OR R2 R2 CS RS CB RB
CF CB = CF || CS RB = R1 || RS R1
RS
CS CB RB
AD745
INVERTING CONNECTION
OUTPUT
AD745
OUTPUT
NONINVERTING CONNECTION
Figure 40. Optional External Components for Balancing Source Impedances
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
16-Lead SOIC (R) Package
0.4133 (10.50) 0.3977 (10.00)
16
9
0.2992 (7.60) 0.2914 (7.40)
1 8
0.4193 (10.65) 0.3937 (10.00)
PIN 1
0.050 (1.27) BSC
0.1043 (2.65) 0.0926 (2.35)
0.0291 (0.74) 0.0098 (0.25)
45
0.0118 (0.30) 0.0040 (0.10)
8 0.0192 (0.49) SEATING 0 0.0125 (0.32) 0.0138 (0.35) PLANE 0.0091 (0.23)
0.0500 (1.27) 0.0157 (0.40)
Revision History
Location Data Sheet changed from REV. C to REV. D. Page
Deleted 8-Lead Plastic Mini-DIP (N) and 8-Lead Cerdip (Q) Packages from CONNECTION DIAGRAM . . . . . . . . . . . . . . . . . . 1 Edits to PRODUCT DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Edits to ELECTRICAL CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Deleted to METALIZATION PHOTOGRAPH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Deleted text from HOW CHIP PACKAGE TYPE AND POWER DISSIPATION AFFECT INPUT BIAS CURRENT . . . . . . . . 9 Deleted 8-Lead Plastic Mini-DIP (N) and 8-Lead Cerdip (Q) Packages from OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . 12
-12-
REV. D
PRINTED IN U.S.A.
C00831-0-3/02(D)
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